6-gang FM stereo tuner heads

[regenfreak]

G0HZU_JMR In the case of the Siglent analyser, I don't think you can use the +15dBm input TOI (of the first mixer) for your calculations unless you use really wide tone spacings, eg 20MHz spacing.

This is because the Siglent appears to have been designed with a different gain distribution to a classic spectrum analyser.

The old HP 8566 and 8568 analysers don't have any amplification until after the second mixer. This means the distortion performance on all but the narrowest spans will be dictated by the first mixer. Your Siglent analyser has a lot of amplification after the first mixer and so the input TOI of the second mixer will probably limit the overall input TOI. I think this means that the effective input TOI will be lower than +15dBm with 0dBm input attenuation. The datasheet suggests that it will be about +10dBm.

The noise floor with a 10kHz RBW at 10dB attenuation is going to be about -90dBm and this is equivalent to -130dBm/Hz.

Your SFDR calculation should be done at the chosen RBW setting.

Cheers. I can try to draw a block diagram of the Siglent too based on EEBlog video too in a day or two.
20M spacing seems massive. I am not sure how you got this estimae. My calculation was based on RBW 10KHz

Normally, when measuring the TOI of an external DUT (rather than the TOI of the analyser itself), it's best to do it at a fairly wide tone spacing if allowed and if possible. The HP 8566 and 8568 only have an average input TOI, so one way to get around this is to space the tones a long way apart. eg 1MHz apart. This avoids having high phase noise interference at the 2f2-f1 and 2f1-f2 IMD tone frequencies.

Then set the analyser to a very narrow span of maybe 5kHz so a narrow RBW can be used and then visit 2f1-f2, f1, f2 and 2f2-f1 in turn to measure the amplitude and then work out the TOI.

This method has the advantage of avoiding the worst of the analyser's phase noise and it also allows a very narrow RBW. This means the analyser can be run with a higher input attenuation setting and this helps with the overall measurement uncertainty of the DUT as it improves the input TOI of the HP 8566/8 analyser.

I knew it is a double edged saw. The wide spacing measurement has poorer DANL The narrow spacing measurement has higher phase noise contamination.
Now do you take the worst case out of the two TOI values? Or take the average?

The isolation will be about 10dB worse. However, in the past, I've fitted an RC network followed by a 10dB attenuator at the sum port of the TSC-2-1 and optimised the RC values for max isolation for a chosen band. I've done this at the popular satellite IF frequencies of 70MHz and 140MHz in the past. With the RC network suitably tweaked, it's possible to get a very high isolation over a very narrow bandwidth, maybe across a few MHz or so.

The measurement of TOI of a DUT becomes increasingly like a SAS obstacle course. Ultimately, there must be a middle ground, otherwise the strict, enveloping-pushing requirements on spectrum analyzer, combiner, signal sources and diplexers are so demanding that it means total surrender; ok dude, I will just take the TOI from the manufacturer's datasheet. Having said that understanding how the spectrum analyzer works internally is rewarding and intellectually challenging.

The inside of spectrum analyzer is like a magic box to be explored. I am losing interest in Ham radio. I must confess that I find VNA and Spectrum analyzer on their own are much more interesting than Ham radios.

I've got loads of TSC-2-1 combiners here in a tube and I'll see if I can do this again at 100MHz. It might not be until the weekend though.

Thanks.

[G0HZU_JMR]

20M spacing seems massive. I am not sure how you got this estimate.

It was a guess because I don't know the bandwidth of the first IF filter in the Siglent analyser. I think it will be less than 20MHz but probably not by much. If the tones are spaced at a wider spacing than the bandwidth of the first IF filter in the Siglent analyser, then the second mixer will not see both tones at the same time. Therefore, the first mixer and the IF amplifier after it will dictate the input TOI.

This might improve the input TOI, but a lot depends on how well the IF1 amp performs when feeding the test tones into the stopbands of the IF1 filter that follows the IF1 amplifier. All this stuff is hard to judge, but the gain distribution in the Siglent analyser does look a bit unconventional to me.

The HP 8568 analyser overcomes this issue by using an attenuator at the IF port of the mixer (to improve the IF port termination here) and the first stage of amplification is after the second mixer. The HP 8566 uses a directional filter (a bit like a diplexer) in place of the 6dB attenuator and this costs more and should perform better.

This gives a consistent input TOI right through the first two mixer stages. That's how spectrum analysers were designed 40-50 years ago. Modern analysers do now tend to have an amplifier after the first mixer, but it typically has a low gain and a very high input TOI. This IF1 amplifier stage improves the DANL by a few dB for one thing. It can also introduce some much-needed reverse isolation between the first two mixers. The IF1 amplifier in the Siglent looks to have >15dB gain at the IF1 frequency and this seems very high to me.

[G0HZU_JMR]

The measurement of TOI of a DUT becomes increasingly like a SAS obstacle course. Ultimately, there must be a middle ground, otherwise the strict, enveloping-pushing requirements on spectrum analyzer, combiner, signal sources and diplexers are so demanding that it means total surrender

It's often an iterative process, I start out with the basic setup I showed earlier and then tweak it as required. I've got lots of different combiner types here, lots of filters and lots of attenuators including 1dB step attenuators. Some sig gens are better than others. I prefer to use a generator that can turn the ALC bandwidth right down.

The analyser's internal IMD should ideally be 20dB lower than the DUT and this can make it harder to achieve a working setup if trying to measure DUT IMD at -75dBc for example. The analyser IMD (and the sig gen and combiner IMD) has to be lower than -95dBc in this case.

[regenfreak]

It is probably time consuming to draw the blocks. Just quickie, so far I have:

From input:
1. GaAs switch TOI = +45dm (unknown chip, just based on similar)
2. pre-amp unknown IC ((have to dig further)
3. GaAs switch TOI = +50dbm HMC284AM,
4. 20dn attenuator TOI = +44dbm, HMC307
5. LP
6. 1st mixer HMC488 3.1-3.9G, TOI = +15dbm
7. 18db amplifer HMC716 NF = 1.0, TOI = +33dm
8. BF/LPF
9. 2nd mixer HMC488? 3.1-3.9G, TOI = +15dbm
10. 810 MHz saw filter NDF 8027
11. 3rd Mixer ADE-2 mixer TOI = +20dbm

I downloaded the hi res photos in the descriptions of the EEVbllog video so I may try to identify the unknowns.

[Radio Wrangler]

Hittite (ADI subsidiary nowadays) do indeed have some nice parts, but there are also some others hidden in Skyworks' catalogue.

Oh, a lot of SAW filters have spurious responses, so may need another frm of filter to cover for them.

David

[G0HZU_JMR]

I just had another look at the Siglent teardown, and I think the first IF filter will be a lot, lot wider than 20MHz. The filter is just a low cost printed interdigital filter. The finger spacing is quite close so this filter could be several hundred MHz wide.

[regenfreak]

i see. The first GaAs switch is MASWSS0181TR-3000, TOI = +57 dbm

I have found this TOI test methods created by two Ham operators using just one sig gen at 3 and 4MHz( Siglent application notes):

https://www.siglenteu.com/applicatio...ting/?pdf=8970

[G0HZU_JMR]

Yes, there are ways to extend the range of the test gear. Using BPF and BRF can work well.

Luckily for me, I rarely have to measure IMD terms that are required to be lower than about -80dBc. When I need to measure IMD this low, this is usually done at the IF output port of a downconverter, and this can be in the range of maybe 20MHz to 300MHz. Usually, it's in the range of 50MHz to 150MHz. I've been doing this stuff for decades and in my experience, it's fairly easy to achieve -100dBc IMD from the sig gens and combiner when using the simple test setup I showed earlier. This is all I need in order to measure -80dBc IMD with low uncertainty. The tool of choice at work for the last 20 years has been the Agilent E4440A PSA analyser and in this frequency range it has 104dB SFDR with 30Hz RBW. I'm lucky in that I've also got one of these analysers here at home.

I've never seen an issue when using a 6dB hybrid combiner when trying for -100dBc IMD in this frequency range and this is when producing tones at up to 0dBm each. It might be different if I was trying for tones at (say) 13dBm each. So far, I've never needed to do critical IMD testing with tone levels that high. I'm not sure what the TSC-2-1 can deliver in terms of IMD free test tones, but I'd expect it to be in the ballpark of -100dBc at maybe -10dBm per tone. The last time I played with a TSC-2-1 at 70MHz or 140MHz was about 30 years ago and -72dBc IMD was our usual design requirement in those days.

[Radio Wrangler]

When having to do fairly extreme TOI measurements in the past, i've sometimes been forced into using older valve-based sig gens with tuned power amp output stages to get broadband noise down. I was suffering from noise*noise convolution with noise and phase noise in the DUT (a receiver) The DUT was giving a floor below -140dBm with 22Hz FDM pilot filter selected.

David

[regenfreak]

Originally Posted by Radio Wrangler

When having to do fairly extreme TOI measurements in the past, i've sometimes been forced into using older valve-based sig gens with tuned power amp output stages to get broadband noise down. I was suffering from noise*noise convolution with noise and phase noise in the DUT (a receiver) The DUT was giving a floor below -140dBm with 22Hz FDM pilot filter selected.

David

You are saying that valve signal generator is having lower thermal noise than solid state counterparts. The phase noise is proportional to frequency of a negative power nth except the PLL and broadband noise components are independent of frequency (attachment). If OcXO or TCXO are used, the main contributor of phase noise near the carrier, the reference frequency would be reduced.

It is not easy to build my own frequency crystal oscillator source. It must have very low phase noise and 50 ohms output. I have no idea at the moment.

I attached the measurement of phase noise of a crystal oscillator. It seems insanely difficult to get reliable results. You multiply a square wave with an anti-phase sine wave from the crystal oscillator to eliminate the carrier, obviously. You make a rabbit disappearing in your hat. The product of the two frequency sources is the difference, giving the noises of the sig gen only after some filtering. So far so good until you know the mixer is the weakest link that it is not really a simple Math multiplication operation of a square wave and a sine wave. What about the noises of the square source? So the measurement is plagued with flaws.

he tool of choice at work for the last 20 years has been the Agilent E4440A PSA analyser and in this frequency range it has 104dB SFDR with 30Hz RBW. I'm lucky in that I've also got one of these analysers here at home.

I have been reading PSA application notes since last weekend and where i got the equations from in my calculation. Thats when things start to make more sense when they elaborate their points in more details. The optimum TOI mixer level is about -38dbm for the E440A at band 0.

[G0HZU_JMR]

I attached the measurement of phase noise of a crystal oscillator. It seems insanely difficult to get reliable results.

There are commercial phase noise analysers available that use similar techniques to measure phase noise.

If it helps, I had a go at making a low cost, low noise crystal oscillator last year. I used a class A amplifier in the loop and used classic theory to predict the close in phase noise. See the simulated phase noise and the actual phase noise as measured by a phase noise analyser at work. The spikes/spurs on the trace are from nearby equipment in the lab at work. Even with the lab interference, the result looks to be very close to the theoretical result. I tried to design an oscillator that could get close to the limits of the phase noise analyser. This meant driving the crystal quite hard.

The other phase noise plot is of a 102.4MHz VCXO. This VCXO is in a PLL and is locked to a 10MHz OCXO. This is a very low noise VCXO and it was quite expensive. A low-cost 102.4MHz oscillator would probably be 10-15dB noisier.

You can buy a basic 100MHz crystal oscillator from Mouser or Farnell for about £3 although I'd expect the phase noise to be quite a bit higher than the VCXO. It would still be good enough for IMD testing though.

[regenfreak]

Originally Posted by G0HZU_JMR

There are commercial phase noise analysers available that use similar techniques to measure phase noise.

If it helps, I had a go at making a low cost, low noise crystal oscillator last year. I used a class A amplifier in the loop and used classic theory to predict the close in phase noise. See the simulated phase noise and the actual phase noise as measured by a phase noise analyser at work. The spikes/spurs on the trace are from nearby equipment in the lab at work. Even with the lab interference, the result looks to be very close to the theoretical result. I tried to design an oscillator that could get close to the limits of the phase noise analyser. This meant driving the crystal quite hard.

The other phase noise plot is of a 102.4MHz VCXO. This VCXO is in a PLL and is locked to a 10MHz OCXO. This is a very low noise VCXO and it was quite expensive. A low-cost 102.4MHz oscillator would probably be 10-15dB noisier.

You can buy a basic 100MHz crystal oscillator from Mouser or Farnell for about £3 although I'd expect the phase noise to be quite a bit higher than the VCXO. It would still be good enough for IMD testing though.

Cheers. I ordered a cheapish 100MHz TCXO from china and it is on its way. Hope I dont need a PLL for that.

The TOI techniques by the two Hams AA7U and N4CY in Siglent application notes (post #307) is the way to go for me. What they did was clever.

Instead of measuring the fundamental and third order distortion products simultaneously, they split the test into two parts. This addresses the tug-of-war between opposing constraints; phase noise due to tone spacing and noise floor. The band-stop filter eliminates the fundamentals before the low-level modulation products going into the spectrum analyzer, making the mixer level of the spectrum analyzer less of an issue. No hooligan can enter the stadium at the gate. We can measure the distortion products with very small RBW and then add the insertion losses of the band stop filter. They used classic bandpass filters for the output ports of a single signal generator and it seems to be effective. Since the Chebyshev has rather narrow bandwidth and steep skirts, the reflected energy of the harmonics back to the signal generator are not escaping. Even you can argue there could be cross-talk leakage two channels of the spectrum analyzer. Their results show it is not the case.

It is easy to draw comparison between hobbyist grade Siglent Spectrum analyzers to HP pro classic workhorse. But they are apple and orange, and their price differences are over an order of magnitude. Secondly, I dont do this for a living in a test lab, no boss tells me off if my measurement is below industrial standard, so we can do whatever we got with limited budget and equipment. Otherwise if it is so much hassle to do a simple TOI measurement, I would rather go to Youtube watching some cat videos...

BTW their resistive-transformer type of power combiner is impressive, close to 78db isolation at 3.5MHz. I have measured my chinese power combiner (attached), it is about 47db port isolation and -3db insertion loss at 10MHz. I am sure it is a hybrid type too. I want to replicate their combiner but can't find any T106-15 toroids for sale in the UK.

I am going to make a 7th order Chebyshev band stop filter with cut-off frequencies at 9MHz and 11 MHz in order to measure the fundamental tones at 10MHz (with a cheapo OCXO) and at 11MHz with a Rigol DG1022Z.

One interesting point they mentioned in the Siglent TOP application notes is that the Siglent spectrum analyzer has a fraction subharmonic spur at 5MHz and they use odd fundamental tones (last two decimal points) in their measurement. I guess the subharmonic spur coming from the frequency multiplier of the spectrum analyzer.

[G0HZU_JMR]

A low-cost way to measure close-in phase noise with your Siglent analyser would be to make a notch filter using crystals. I've done this in the past with very good results. The downside is that the phase noise can only be directly measured at the frequency of the notch filter. However, if a low noise LO and a mixer is used to convert the DUT to the frequency of the notch the other DUT frequencies can be measured using the same notch filter.

The notch below was made using crystals that cost about 20p each. The span is only 10kHz and the notch is 70dB deep. You can see that the notch is very sharp.

This effectively nulls the carrier from the DUT, and this filters away the contribution from the analyser phase noise. The VNA plot below shows the filter response. The other plot shows it in action where three signal sources are compared for close in phase noise. The carrier power is 0dBm, so the marker reading is in dBc/Hz. You can see that the cleanest source was a crystal oscillator made using one of the 20p crystals. It managed -163dBc/Hz at about 5kHz offset. The other two sources were an Agilent vector sig gen (noisy) and an old Marconi 2019 sig gen.

Your Siglent analyser would probably be able to measure down to about -155dBc/Hz with this setup if the preamp was enabled. This is over 50dB better than the analyser can manage without the notch filter.

[G0HZU_JMR]

I am going to make a 7th order Chebyshev band stop filter with cut-off frequencies at 9MHz and 11 MHz in order to measure the fundamental tones at 10MHz (with a cheapo OCXO) and at 11MHz with a Rigol DG1022Z.

There are various ways to do this stuff, but in your case, I'd recommend a different approach.

If I assume you want to do IMD measurements at the 10.7MHz IF of a VHF FM receiver, then it would be wise to do this within the 250kHz bandwidth of a typical 10.7MHz ceramic filter.

One way to do this is with the setup below. The 10.7MHz SSB filter will only be a few kHz wide, and the idea would be to tune the LO (or the test tones) to centre 2f1-f2, f1, f2 and 2f2-f1 at 10.7MHz in turn and then you can compute the TOI.

The setup below would be immune to analyser phase noise effects and also analyser IMD limitations assuming the tone spacing was set at (say) 50kHz.

[G0HZU_JMR]

Actually, because it's FM you probably only need to do adjacent channel IMD testing, so the spacing can be a bit wider. Realistically, the tones and the IMD would need to be close enough to go through the (1MHz wide?) tracking preselector to get to the mixer.

Your Siglent analyser can still achieve quite a good SFDR in a 10Hz RBW even when phase noise is factored in. It's probably going to be fine for testing each of the various input stages of a VHF FM receiver assuming that you are going to be using something like a level 7 diode ring mixer or a MOSFET mixer.

[regenfreak]

Originally Posted by G0HZU_JMR

There are various ways to do this stuff, but in your case, I'd recommend a different approach.

If I assume you want to do IMD measurements at the 10.7MHz IF of a VHF FM receiver, then it would be wise to do this within the 250kHz bandwidth of a typical 10.7MHz ceramic filter.

One way to do this is with the setup below. The 10.7MHz SSB filter will only be a few kHz wide, and the idea would be to tune the LO (or the test tones) to centre 2f1-f2, f1, f2 and 2f2-f1 at 10.7MHz in turn and then you can compute the TOI.

The setup below would be immune to analyser phase noise effects and also analyser IMD limitations assuming the tone spacing was set at (say) 50kHz.

Thanks. Sounds like a good idea. Please correct me if I misunderstand your idea:

Take tone spacing of 50khz, f1= 10.65, f2=10.75MHz. You measure the f1, f1, 2f1-f2, 2f2-f1 in four steps sequentially,

For high sided LO as an example:

2f2-f1 thru measurement:
LO-(2f2-f1) = 10.7
LO = 10.7 + 2f2-f1= 10.7 + 2x 10.75-10.65=21.55MHz

f2 thru measurement:
f2-LO= 10.7
LO=10.75-10.7=50kHz

Similarly for the rest.

For low sided LO:
(2f2-f1)-LO = 10.7
LO = 2f2-f1 -10.7=2x 10.75-10.65-10.7= 150khz etc....

I hope i got the Maths right..I am in a hurry and is at work during lunch break.

A low-cost way to measure close-in phase noise with your Siglent analyser would be to make a notch filter using crystals. I've done this in the past with very good results. The downside is that the phase noise can only be directly measured at the frequency of the notch filter. However, if a low noise LO and a mixer is used to convert the DUT to the frequency of the notch the other DUT frequencies can be measured using the same notch filter.

The notch below was made using crystals that cost about 20p each. The span is only 10kHz and the notch is 70dB deep. You can see that the notch is very sharp.

This effectively nulls the carrier from the DUT, and this filters away the contribution from the analyser phase noise. The VNA plot below shows the filter response. The other plot shows it in action where three signal sources are compared for close in phase noise. The carrier power is 0dBm, so the marker reading is in dBc/Hz. You can see that the cleanest source was a crystal oscillator made using one of the 20p crystals. It managed -163dBc/Hz at about 5kHz offset. The other two sources were an Agilent vector sig gen (noisy) and an old Marconi 2019 sig gen.

It is another neat idea. This adds to my to-do list. I have not taken much interest in phase noise measurement. I am wary of using a mixer in phase noise measurement as the noises of two other sources add to the uncertainties. What do you think of the Marconi 2019 sig gen?

Actually, because it's FM you probably only need to do adjacent channel IMD testing, so the spacing can be a bit wider. Realistically, the tones and the IMD would need to be close enough to go through the (1MHz wide?) tracking preselector to get to the mixer.

Your Siglent analyser can still achieve quite a good SFDR in a 10Hz RBW even when phase noise is factored in. It's probably going to be fine for testing each of the various input stages of a VHF FM receiver assuming that you are going to be using something like a level 7 diode ring mixer or a MOSFET mixer.

I am planning to build the third FM tuner using the 6-gang air variable capacitor from the UK Ministry of Defense. I have three of them, NOS mint in shiny silver with manufacturers' boxes. Originally, I wanted to use double balanced DGMs for the mixer. But it would need a IF transformer with primary tapping connected to the drains and sources. I am not sure how it works in terms of phasing and impedance matching...etc. So I may opt for single FET mixer design. It is easy to implement diode ring mixers in 50 ohm homebrew QRP receivers but it is hard to get impedance matching right in FM broadcast front end. I dont have the brain power to do the design calculations.

One of the biggest headaches is none of FM broadcast tuners use 50-ohm circuits, they are full of high impedance nodes at both I/O stages. Unlike the QRP, CW and SSB homebrew projects, there are zillions of schematics online that use 50 ohm topology, making the impedance matching less of an issue. If you use a matching pad for the FM antenna input and stick a hi-Z FET probe into different hi-Z nodes of the RF stages or mixer, it is difficult to isolate the effects of impedance mismatch and the FET probe's contribution of TOI measurement, i.e. it will get muddy fast. I have not thought about adjacent channel IMD testing. I dont know how manufacturers could get around the 50ohm-hi-Z mismatch issues in the 1970s, they could have some form of dedicated broadcast receiver analyzers to do the job.

HP made spectrum analyzers with 1M ohms impedance input...

[regenfreak]

One way to do this is with the setup below. The 10.7MHz SSB filter will only be a few kHz wide

I would have to find a narrowband crystal with known datasheet, having values of Rp and Cp. Applying transformation to find Rs and Cs, the LC matching pads can be found by using the Smith chart.

The HP spectrum analyzers such as the HP 3585A uses 1M ohms impedance transformation buffer amp at the input with a impedance selector switch. I suppose the buffer amp would improve the signal-to-noise ratio but also add to the TOI contribution in the chain.

[Radio Wrangler]

Spectrum analyser crystal filter design is somewhat contorted to making programmable bandwidths and synchronous tuning shape (not actually Gaussian, though close enough up top)

David Ford's gang did the 3585A over at Loveland, but he'd been at the 'ferry for some time. I think my write-up of the simpler spectrum analyser filter set (in the 3724A) is in the HPJ article (I no longer look like a hairier Harry Potter).

The higher freq analysers use PIN diodes to control stage Q and hence make variable bandwidth.

With so many resistors floating around, loss == noise floor takes a hit.

David

[G0HZU_JMR]

Thanks. Sounds like a good idea. Please correct me if I misunderstand your idea:
Take tone spacing of 50khz, f1= 10.65, f2=10.75MHz. You measure the f1, f1, 2f1-f2, 2f2-f1 in four steps sequentially,

That's the idea although I spotted that you have 100kHz tone spacing in your example. That's still fine as long as you work out the correct frequencies to measure in each case.

What do you think of the Marconi 2019 sig gen?

I've got two of them here and I've been using 2019 sig gens since the 1980s at work.

Sadly, these generators have reliability issues. They don't have any fan cooling and it's easy to block the cooling vents underneath the sig gen. They then tend to run a bit hot, and this shortens the life of the electrolytic caps inside. The PSU and transformer are weak points in the design and can fail.

When the caps start failing the sig gen will produce quite a few low-level spurious signals on the output. The output mechanical step attenuator can get sticky and intermittent if not used regularly. For these reasons this sig gen probably isn't a good purchase in 2022 unless it has been serviced or the price is very low, and you don't mind doing the work yourself.

The RF performance is quite good, with low phase noise across the 2.5-50MHz range. Above about 520MHz it uses a doubler and the phase noise is relatively poor above 520MHz. The RF levelling accuracy is very good up to about 520MHz. Above this the levelling is fairly lumpy and it gets worse above 900MHz.

[regenfreak]

For me, it is nothing short of magic to create a 1Hz bandwidth RBW filter using crystals before the introduction of digital IF.

Up to this point, i have not paid much attention to crystals as they fall into the territories of the QRP homebrewers who want to make their own narrow bandwidth CW or SSB filters. I haven't got time to play with the measurement of crystal parameters, ladder filter, Dishal design programme, crystal matching and all the jazz..One can spend ages trying to understand the theory, designing, buying hundred of crystals and sorting them, another black hole to walk into. Everything is revolving around Amateur radio and i am at the peripheral, trying not get sucked into it. I dont find the buccal speech, gibberish exchanges and constant battle with RFI in Ham radio appealing.

Sadly, these generators have reliability issues. They don't have any fan cooling and it's easy to block the cooling vents underneath the sig gen. They then tend to run a bit hot, and this shortens the life of the electrolytic caps inside. The PSU and transformer are weak points in the design and can fail.

When the caps start failing the sig gen will produce quite a few low-level spurious signals on the output. The output mechanical step attenuator can get sticky and intermittent if not used regularly. For these reasons this sig gen probably isn't a good purchase in 2022 unless it has been serviced or the price is very low, and you don't mind doing the work yourself.

The RF performance is quite good, with low phase noise across the 2.5-50MHz range. Above about 520MHz it uses a doubler and the phase noise is relatively poor above 520MHz. The RF levelling accuracy is very good up to about 520MHz.

We are posting at the same time. Oh well. I see them on ebay from time to time. So there is always a risk that they may fail.

That's the idea although I spotted that you have 100kHz tone spacing in your example. That's still fine as long as you work out the correct frequencies to measure in each case.

cheers. I have only noticed it now. I did it in a hurry writing the calculations in a napkin over lunch.